PFC circuits with very low THD

ABSTRACT

A boost chopper circuit is described that an alternating current (AC) power source; at least one inductor connected to said AC power source; a rectifier connected to said inductor and AC power source; at least one switch shorting our said rectifier; a series circuit connected in parallel with said switch of at least one diode and a capacitor; and a load connected in parallel with said capacitor. A control technique is employed that includes turning on and off the switch in order to keep the average current per pulse cycle proportional to the AC input voltage during the same pulse cycle.

CROSS-REFERENCE TO RELATED APPLICATION

The present application is a divisional of U.S. application Ser. No.15/588,563, filed May 5, 2017, which is a divisional of U.S. applicationSer. No. 14/536,580, filed Nov. 7, 2014, which claims the benefit ofprovisional application No. 61/902,121, filed on Nov. 8, 2013, entitledPFC CIRCUITS WITH VERY LOW THD, which was incorporated by reference.

FIELD

Embodiments described herein relate to power factor correction (“PFC”)circuits; and more specifically, to PFC circuits with very low totalharmonic distortion (“THD”).

BACKGROUND

THD (Total Harmonic Distortion) and the Power Factor (pf) in 1-phase and3-phase Alternating Circuit (AC) input lines have been major problemsfor many years. The THD of 1-phase and 3-phase systems withInductor-Capacitor (LC) filters is 120% and 28%, respectively. Withcapacitor input filters, the THD is considerably higher.

Regulatory authorities continue to increase their specifications. Whenthey learn details of the present embodiments, it is possible they maywant to significantly increase their specifications.

SUMMARY

Embodiments described herein can obtain a pf=1 and a THD of less than0.001% when minimum cost is not a requirement.

The described embodiments include two different methods that use threedifferent basic topologies to obtain a very low THD (<=0.01%) and a pf=1in 1-phase and 3-phase input voltage systems.

The first method includes a control method to vary the pulse width of agate driver in such a way that the average current during the pulsecycle is proportional to the AC voltage during that pulse cycle. Thiscontrol method is done each pulse cycle.

The second method includes using a concept wherein the average currentduring the pulse cycle is naturally proportional to the AC voltageduring each pulse cycle using two different basic topologies. The twobasic topologies are using half bridges and full bridges in theiroutputs.

The full bridge versions are best suited for fixed voltage loads. Thehalf bridge versions are best suited for variable voltage outputs eithercurrent or voltage regulated.

Other objects, features, and advantages of the embodiments describedherein will be apparent from the accompanying drawings and from thedetailed description that follows below.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments described herein are illustrated by way of example and notlimitation in the figures of the accompanying drawings, in which likereferences indicate similar elements.

FIG. 1A is a prior art drawing of a boost chopper.

FIGS. 1B-1E are used in detailing a problem of FIG. 1A.

FIGS. 1F-1G are used in detailing one of several control concepts thatsolves the problem of FIG. 1A.

FIG. 2A is the same drawing as FIG. 1A.

FIGS. 2B-2D illustrate details of several control concepts that solvethe problem of FIG. 1A.

FIG. 3A is a schematic of a circuit where the average current per pulseis proportional to the AC voltage present at the time of the pulse. Nopulse width modification is required to get a low THD.

FIGS. 3B-3D are used to describe the operation and design of FIG. 3A.

FIG. 4 is the same as FIG. 3A where the bi-directional switch of FIG. 3Ais replaced with four diodes and one switch.

FIG. 5A is the same as FIG. 4 with the transformer moved in series withthe switch (point Z of FIG. 4). Note that because the voltage applied tothe transformer is only positive, only a half bridge is required in thesecondary.

FIG. 5B is a modification of FIG. 5A wherein two primaries of thetransformer are used. When the AC voltage reverses polarity, the otherprimary is used.

FIG. 6 is a 3-phase version of FIG. 5B. Two different shorting circuitsare shown shorting out the output of the 3-phase bridge.

FIGS. 7A-7D are other 3-phase versions of FIG. 5B.

FIG. 7E is a non-isolated buck-boost circuit with the same low THD asthe other circuits shown.

FIGS. 7F-7G are other versions of FIG. 1A, which is prior art.

DETAILED DESCRIPTION

The new topologies described below are low cost AC to DC power factorcorrection (PFC) circuits that have very low Total Harmonic Distortion(THD) in the AC input that are very cost effective from watts tomegawatts. The new topologies work in both 1 and 3-phase AC lines andrequire only one switching device. Most of the new topologies can be runeither isolated or non-isolated with negligible difference in cost. Ithas been found that a standard boost circuit has a high THD except undersome extreme conditions. One of the embodiments is a method ofcontrolling the pulse width of the control signal of a standard boostcircuit in such a way that the THD is lowered to a very low value—i.e.,<0.1%, under most conditions. The power factor (pf) in the AC lines isone. The normal THD in 1 and 3-phase systems with an inductive inputfilter is approximately 120% in single phase and 28% in 3-phase systems.In 3-phase applications, several different topologies have been foundthat give the same low THD and high pf while still using only oneswitching device. In some topologies it is advantageous to use twoswitching devices with lower voltage rating and lower switching lossesthan the one switching device topologies. The new topologies can be runeither isolated or non-isolated with negligible difference in cost.

Shown in FIG. 1A is an alternating current (“AC”) to direct current(“DC”) boost chopper 100, which is prior art. There are many differentconfigurations of circuitry of boost choppers that obtain the same endas boost chopper 100. For example, the inductor (L) 102 can be put atpoint 104 of FIG. 1A. Shown in FIG. 7F and FIG. 7G are two otherversions. The boost chopper 100 takes the AC input voltage, V_(t), andconverts it to a DC output voltage (V_(O)). The gate drive to theinsulated-gate bipolar transistor (IGBT) 106 runs at a fixed frequencyand fixed pulse width (under constant load and input AC voltageconditions). The circuit of the boost chopper circuit 100 runs in whatis known as discontinuous mode—i.e., the gate pulse to the IGBT 106 isnot turned on again, after being turned off, until the current in L 102(I_(L)) has gone back to zero current. The output voltage has to behigher than the highest peak voltage, V_(P), of the input voltage,V_(t). The output voltage V_(O) is controlled by the IGBT gate pulsewidth of IGBT 106—i.e., the wider the pulse width the more the outputvoltage (V_(O)) will be for a fixed load.

Shown in FIG. 1B is a graph 199 illustrating the voltage seen betweenpoints A and B of FIG. 1A. Assuming the voltage drop across diodes 108Ato 108D is small, the voltage between point A and point B is expressedas VAB=|VP Sin O—|. Shown in FIG. 1C is a graph 197 that is an expansionof FIG. 1B for a ½ cycle of the input voltage V_(t). Also shown in FIG.1C is the current I_(L) and in dotted line the average value of I_(L).In FIG. 1C, VO is greater than Vp.

Note that the average value of I_(L) has a very low THD (Total HarmonicDistortion). The low THD will be obtained is when V_(O) is much largerthe V_(P). V_(O) has to be at least 30 times the value of V_(P) in orderto have a low THD. The more V_(O) is less than 30 times V_(P), thegreater the THD will be.

It has been found that, in most applications of the circuit shown inFIG. 1A, the output voltage is much less than required for a low THD.Shown in FIG. 1D is a graph 193 illustrating an expansion of the currentshown in FIG. 1C inside section 195 of FIG. 1C. Note that with the highvalue of V_(O) the fall time of the current from its peak value of I_(P)is much less than the rise time. It is this fast fall time that isrequired to give a low THD in the circuit shown in FIG. 1A. When thefall time is fast the average current during any pulse cycle isproportional to the input AC voltage during any pulse cycle andtherefore there is a very low THD in the AC input voltage.

Shown in FIG. 1E is a graph 191 illustrating what the current will looklike during a pulse cycle when the AC voltage is at its peak value, whenthe output voltage (V_(O)) is only three times V_(P). The fall time ofthe current is not very fast compared to the rise time. This causessignificant THD in the input current. The average current I_(L) (duringa pulse cycle) is not proportional to V_(AC). Note that when V_(AB) isnot near the peak value the fall time is faster and faster as V_(AB) isat a lower and lower value.

In order to get a low THD when V_(O) is not many times the V_(P) thegate pulse width has to be shortened so the average current during thepulse cycle is the same as when V_(O) is much greater than V_(P). Shownin FIG. 1F is a graph 189 illustrating what the current I_(L) looks likewhen V_(O) is very large (triangle ABC) and when V_(O) is only 3*V_(P)(triangle AB¹C¹) as depicted in FIG. 1E. In order to get low THD, thearea of triangle ABC has to be equal to the area of triangle AB¹C¹.

The equation 187 for the shorter pulse as shown in FIG. 1G is:

${t\; 3} = {t\; 1*\left( \sqrt{1 - \left( \frac{V_{AB}}{V_{O}} \right)} \right)}$

wherein t₁ is the time the pulse would be on with a very high V_(O) andt₃ is the time required to give the same average current as when V_(O)is very high. Note that this equation has to be calculated for each gatepulse because V_(AB) changes as a function of time. Also note that whenV_(AB) is very small t₃ is essentially equal to t₁. Another version ofthe equation shown above and also shown in FIG. 1G is:

${t\; 3} = {t\; 1*\left( \sqrt{1 - \left( \frac{{V_{P}*{Sin}\underset{\_}{O}}}{V_{O}} \right)} \right)}$

With other versions of the AC to DC boost chopper the equation for theshorter pulse in order to get the average current (during a pulse cycle)to be proportional to the input AC voltage could be different but theconcept is the same as described above.

A less complex equation that closely approximates the above equationsis:

${t\; 3} = {t\; 1*\left( {1 - {0.57\left( \frac{V_{AB}}{V_{O}} \right)}} \right)}$

When VP/VO is less than 0.5, the above equation is fairly accurate. Itis especially accurate under constant load conditions. The 0.57 term canthen be adjusted to minimize the THD. The 0.57 term can range fromapproximately 0.5 to 0.65 depending on the details of the design used.

Shown in FIG. 2A is AC to DC boost chopper 200 that is similar to theboost chopper 100 described above in FIG. 1A. The schematic of AC to DCboost chopper 200 is identical to the schematic provided in FIG. 1A. Thedifferences are the use of several control techniques 210, 230, and 250to turn on and off the IGBT 106 in order to keep the average current perpulse cycle proportional the AC input voltage during the same pulsecycle.

FIG. 2B is the general control technique 210 to keep the average currentper pulse cycle proportional the AC input voltage during each pulsecycle. FIG. 2C is the control technique 250 using the equation

${t\; 3} = {t\; 1*\sqrt{\left( {1 - \left( \frac{VAB}{VO} \right)} \right)}}$to keep the average current per pulse cycle proportional the AC inputvoltage during the same pulse cycle. FIG. 2D is the control technique210 using the equation

${t\; 3} = {t\; 1*\left( {1 - {0.57\left( \frac{VAB}{VO} \right)}} \right)}$to keep the average current per pulse cycle proportional the AC inputvoltage during the same pulse cycle.

Shown in FIG. 3A is an AC to DC boost chopper 300 that is similar to ACto DC boost chopper 200 described above in FIGS. 2A-2D. The circuittakes the AC input voltage, V_(t=)=|VP Sin O—|, and converts it to a DCoutput voltage (V_(O)). The gates that drive to the IGBT's runs at afixed frequency and fixed pulse width (under constant load and input ACvoltage conditions). The circuit shown runs in what is known asdiscontinuous mode—i.e., the gate pulse to the IGBTs is not turned onagain, after being turned off, until the current in the transformer305's secondary has gone back to zero current. X*VO (transformer ratiotimes the output voltage) has to be higher than the highest peakvoltage, V_(P), of the input voltage. The output voltage V_(O) iscontrolled by the IGBT gate pulsewidth—i.e., the wider the pulse widththe more the output voltage will be for a fixed load. In FIG. 3A, thecircuitry allowing for (X*VO) is denoted by box 307.

The circuit shown in FIG. 3A has the advantage of always having a fastfall time when the IGBTs 301 and 303 are turned off as shown in graph399 of FIG. 3B—i.e., when the bi-directional switch 302 is turned offthe I_(L) goes to zero in a very fast time, denoted by t on the x-axisof graph 399, compared to the switching frequency as shown graph 397 inFIG. 31) and therefore the harmonic distortion is very low in the ACinput. Shown in FIG. 3C is the waveform 395 between points A and B inFIG. 3A. Shown in graph 397 of FIG. 3D is an expansion of FIG. 3C forhalf cycle of the input voltage V_(t). Also shown in FIG. 3D is thecurrent I_(L) and in dotted line the average value of I_(L). As can, beseen in FIG. 3D, the THD of the input AC current waveform is very low,under some conditions less than 0.01% at a very low cost.

Note that the THD of 1 and 3-phase systems with LC filters is 120% and28% respectively. With capacitor input filters the THD is considerablyhigher. With the present patent application a THD of less than 0.001%can be obtained when minimum cost is not a requirement.

Shown in FIG. 3B is one pulse cycle of I_(L) when V_(AB) is at its peakvalue V_(P) as shown inside the X area of FIG. 3D. ThedI_(L)/dt=V_(P)/L_(M) where L_(M) is the magnetizing inductance of thetransformer T₁. It has been found that to minimize the switching voltageand current of the circuit of FIG. 3A at the same time the gate pulsefor the IGBT's should be near 50% of the switching period of the IGBT'swhen delivering full power to the load. With the 50% requirement it hasbeen found that:L _(M)=(V _(P))²/(16*P ₀ *f ₀)

Where:

V_(P)=the peak input voltage

P₀=the maximum power to the load

f₀=the switching frequency of the IGBT's

For a complete design, the value of X in FIG. 3A has to such thatX*V₀=>V_(P)

Cutting the value of L_(M) in half allows you to do one of thefollowing:

1. Double the power to the load.

2. Deliver full power to the load with the input AC voltage cut in ½.

Shown in FIG. 4 is a AC to DC boost chopper 400 that is the same as ACto DC boost chopper 300 in FIG. 3A except the bidirectional switch 301has been replaced with four diodes 401A-D and one switch 403. Shown inFIG. 5A is a circuit 425 where a primary of the transformer 309 of AC toDC boost chopper 400 is put in point Z of FIG. 4. Note that transformer307 and 309 are not exactly the same. This is because the voltage isonly positive to the primary of the transformer, so only a half bridgeis necessary in the secondary of transformer 309.

Generally circuits like circuit 400 (with full wave output rectifiers)would be used for fixed voltage outputs. Circuits like circuit 425 (halfwave rectifiers) would be used for variable voltage outputs where theoutput can be either voltage or current controlled or voltage controlledwith current limiting.

Shown in FIG. 5B is circuit 450 that is a more complex version of FIG.5A. Basically the transformer 311 has two primaries. When the inputvoltage reverses polarity the other primary is used. The reason forshowing this more complex version is that it is extendable to 3-phasecircuits.

Shown in FIG. 6 is circuit 600 that is a 3-phase version of circuit 450of FIG. 5B using just one IGBT 403 (with gate 430) for all 3-phases.Circuit 600 includes three-phase AC power source 601: capacitors 620,621, and 622; diodes 650, 651, 652, 670, 671, and 672; primary windings630 of transformer 640; primary windings 631 of transformer 641; primarywindings 632 of transformer 642; secondary winding 690 of transformer640; secondary windings 691 of transformer 641; secondary windings 692of transformer 642; diodes 610, 611, and 612: and capacitor 613. Alsoshown in the same figure in dotted lines is a version using two IGBTs603A and 603B. The two IGBTs 603A or 603B can be used with or withoutthe dotted line between points G and F (GF), which have two switches603A and 603B. In a system where the 3-phase input voltages are exactlythe same and the primary inductance L_(M) in the three transformers isexactly the same there will be no current in the GF lead. With the GFconnection, the 3 phase runs completely independent of each other whenthe 3-phase input voltages are not the same. Another reason for the twoIGBTs 603A and 603B is they can be of lower voltage rating than the oneIGBT 403. The switching losses of IGBTs go down with a lower voltagerating. The two IGBTs 603A and 603B are turned on and off at the sametime but can be run at a higher frequency than the one IGBT 403 versionswhich in turn lowers the cost of the isolating transformers.

Most of the transformers shown in the figures in the present patentapplication can be air-core co-axial designs. This is especially true at3-4 kW and above. Besides the lower cost of air-core transformersanother advantage of co-axial is the leakage inductance is much smallerthan normal transformer design. With lower leakage inductance, theswitching losses are significantly reduced when the IGBT's are turnedoff.

Shown in FIG. 7A through 7E are a series of circuits 700, 710, 720, 730,and 740 showing several different ways of obtaining the same resultsshown in the previous figures. The drawings are just some of the manyvariations that are possible and that still are embodiments of thepresent invention.

Shown in FIG. 7A is a circuit 700 that is 3-phase version of circuit 400of FIG. 4. Circuit 700 includes three-phase AC power source 601;capacitors 820, 821, and 822; diodes 850, 851, 852, 870, 871, and 872;primary windings 830 of transformer 840; primary windings 831 oftransformer 841; primary windings 832 of transformer 842; secondarywindings 890 of transformer 840; secondary windings 891 of transformer841; secondary windings 892 of transformer 842; diodes 810, 811, 812,813, 814, and 815; and capacitor 713. The circuit shown shows threedifferent methods of shorting the output of the 3-phase rectifier. Thefirst shown is just a single switch 701 across the output, switch 701having gate 702. The second (shown in box 703) is a method ofeliminating the crosstalk between the 3-phases when input 3-phasevoltages are not equal or if the primary inductance of the threetransformers are not exactly the same.

The third (shown in box 705) is two series connected switches. Thereason for the two series connected switches is that the voltage ratingof the two switches can be less than the other two shorted methodsshown. This can be important when switching losses become an issue. Thelower the voltage rating of an IGBT the lower the switching losses are.The switches shown in box B are normally turned on and off at the sametime.

Note that the current in the dotted lead connected to the common pointof the three capacitors will be zero when the 3-phase voltages are equaland the primary inductance of the three transformers are exactly thesame. With the full wave rectification in the output of the circuitshown in FIG. 7A, the circuit is best suited for fixed voltage outputsas opposed to FIG. 6, which has half wave rectification is well suitedfor variable voltage applications.

Shown in FIG. 7B is a circuit 710 that is another version of circuit 700of FIG. 7A where the 3-phase bridge and shorting circuit across theoutput of the bridge is replaced with three series connected IGBT anddiodes. Note that when all three IGBT's are on you get the same shortingaction as obtained in FIG. 7A.

Shown in FIG. 7C is a circuit 720 that is a modification of circuit 710of FIG. 7B where you can get half wave rectification in the three outputstages that are denoted by boxes 723, 725, and 727. Circuit 720 includesthree full transformers t₁, t₂, and t₃, each with a primary and asecondary.

Shown in FIG. 7D is a circuit 730 that is a modification of a 3-phaseversion of circuit 425 of FIG. 5A. Note that most, if not all, of thesingle phase versions in the present patent can be used for 3-phaseapplications.

Shown in FIG. 7E is a circuit 740 that is a single phase version ofcircuit 730 of FIG. 7D. Circuit 740 is commonly named a buck-boostcircuit. Note that this circuit has no isolation. A buck-boost circuitcan deliver full power to the output above and below the input voltage.If the circuit shown output is rated low voltage high current then theswitching losses in switch 777 will be high due to the high current inthe switch 777. If the circuit shown output is rated a high voltage lowcurrent then the switching losses in switch 777 will be high due to thehigh switching voltage.

In order to optimize the switching losses in switch 777 for both highvoltage and high current operation the following modifications can bemade. If running high current and low voltage operation the diode 775Acan be moved down the inductor as shown where diode 775B is connected.If running high voltage operation, an over wind can be put on theinductor and diode 775A can be moved to where diode 775C is shown. Alsonote that in 3-phase applications the circuit of FIG. 7E diode 775D hasto be added in the lead labeled PQ.

The embodiments disclosed cover all the single and 3-phase isolated andnon-isolated versions (with the full bridge and half bridge outputs)where the current has a natural fast fall time compared to the risetime.

Shown in FIG. 7F and FIG. 7G are two other versions of the prior artfigure shown in FIG. 1A. Specifically, FIG. 7F shows a prior art circuit199 and FIG. 7G shows a prior art circuit 197. Several other versionsare possible.

In the foregoing specification, the embodiments described herein havebeen described with reference to specific exemplary embodiments thereof.It will, however, be evident that various modifications and changes maybe made thereto without departing from the broader spirit and scope ofthe invention. The specification and drawings are, accordingly, to beregarded in an illustrative rather than a restrictive sense.

What is claimed is:
 1. A power factor correction circuit to operate in adiscontinuous current mode comprising: a three-phase alternating current(AC) power source; three capacitors star-connected across thethree-phase AC power source; a first transformer; a second transformer;a third transformer; respective center taps of primary windings of therespective first, second, and third transformers connected to receivethree-phase AC power from the three-phase AC power source; anodes ofthree first diodes connected to respective first ends of thecenter-tapped primary windings of the respective first, second, andthird transformers; cathodes of the three first diodes shorted togetherand denoted as a first short; cathodes of three second diodes connectedto respective second ends of the center-tapped primary windings of thefirst, second, and third transformers; anodes of the three second diodesshorted together and denoted as a second short; a switching deviceconnected between the first short and the second short; respectivesecondary windings of the first, second, and third transformers; anodesof three third diodes connected to the respective secondary windings ofthe first, second, and third transformers; cathodes of the three thirddiodes shorted together as a third short; respective other sides of thesecondary windings of the respective first, second, and thirdtransformers shorted together as a fourth short; a load capacitorconnected between the third short and the fourth short; wherein widthsof fixed frequency pulses applied to a gate of the switching device toturn on and off the switching device are varied to control a loadvoltage across the load capacitor.
 2. The circuit of claim 1, whereinonly one phase of alternating current (AC) power source is used.
 3. Thecircuit of claim 1, further comprising two series-connected switchingdevices.
 4. The circuit of claim 3, wherein fixed frequency pulses areapplied to the two series-connected switching devices.
 5. The circuit ofclaim 3, wherein a common connection point of the two series connectedswitching devices is connected to a common point of the threestar-connected capacitors connected to the three-phase power source. 6.The circuit of claim 1, wherein the first, second and third transformerscomprise air-core co-axial transformers.
 7. The circuit of claim 2,wherein the first, second and third transformers comprise air-coreco-axial transformers.
 8. The circuit of claim 3, wherein the first,second and third transformers comprise air-core co-axial transformers.9. The circuit of claim 4, wherein the first, second and thirdtransformers comprise air-core co-axial transformers.
 10. A power factorcorrection circuit to operate in a discontinuous current modecomprising: a three-phase alternating current (AC) power source; threecapacitors star-connected across the three-phase AC power source; afirst transformer; a second transformer; a third transformer; firstrespective ends of primary windings of the first, second, and thirdtransformers connected to receive three-phase power from the three-phaseAC power source; second respective ends of the primary windings of thefirst, second, and third transformers connected to respective inputs ofa three-phase bridge; a switching device in parallel with an output ofthe three-phase bridge; secondary windings of the first, second, andthird transformers coupled to respective full-wave center-tappedrectifiers; respective outputs of the three full-wave center-tappedrectifiers connected in parallel; a load capacitor in parallel with theoutputs of the three full-wave center-tapped rectifiers; wherein widthsof fixed frequency pulses applied to a gate of the switching device toturn on and off the switching device are varied to control a loadvoltage across the load capacitor.
 11. The circuit of claim 10, whereinonly one phase of the AC power source is used.
 12. The circuit of claim10, further comprising two switching devices that are series connected.13. The circuit of claim 12, wherein a common connection point of thetwo series-connected switching devices is connected to a common point ofthe star-connected capacitors connected to the three-phase AC powersource.
 14. The circuit of claim 10, wherein the three transformerscomprise air-core co-axial transformers.
 15. The circuit of claim 11,wherein the first, second, and third transformers comprise air-coreco-axial transformers.
 16. The circuit of claim 12, wherein the first,second, and third transformers comprise air-core co-axial transformers.17. The circuit of claim 13, wherein the first, second, and thirdtransformers comprise air-core co-axial transformers.